Integrated circuit and method of soft thermal shut down for power devices

ABSTRACT

A thermal control circuit for an integrated power transistor includes a current generator controlled by a turn on signal, a sensing resistance in series with the power transistor, and a current limiter acting when the voltage drop on the sensing resistance overcomes a certain value. The circuit also includes a current amplifier coupled to the output node of the controlled current generator for outputting a drive current that is injected onto a control node of the power transistor. A soft thermal shut down circuit is provided having a conduction state which is enhanced as the temperature increases for reducing the drive current. The circuit controls the voltage on the power transistor in a more effective manner because the current amplifier has a variable gain controlled by the state of conduction of the soft thermal shut down circuit.

FIELD OF THE INVENTION

[0001] The present invention relates in general to thermal control circuits, and, more precisely, to circuits for thermally controlling integrated power devices (smart-power devices) such as driving inductive loads.

BACKGROUND OF THE INVENTION

[0002] Power stages driving inductive loads require, in certain applications, thermal control for gradually decreasing the current when the temperature increases (soft thermal shut down) and until a steady state condition is reached. At steady state, the power dissipated by the Joule effect equals the rate of heat dissipated in the surroundings.

[0003] For example, this kind of thermal control is typically required in integrated power devices employed in electronic ignition systems for engines of motor vehicles. In these applications a soft thermal shut down control circuit avoids the occurrence of an abrupt turning off of the power device in series with the coil upon reaching an abnormally high temperature because of possible anomalous functioning. Generation of sparks at the plugs and undesired detonations are thus prevented.

[0004] A typical prior art circuit that performs such a function as depicted in FIG. 1A includes functional elements that play a role during an ON phase of the power stage. Instead of a single bipolar transistor, several transistors in a Darlington configuration, or even in a three-stage configuration, if the current absorption by control circuits must be small, may be used.

[0005] When a turn on signal “IN” switches from a low logic state to a high logic state, the current generator Iref is enabled. This generator, through a current amplifier A1 _(I) with a constant gain A1, turns on the power stage by forcing a drive current Ib. In these conditions, the collector current Ic of the power stage increases and its variation depends on the value of the inductance of the load and on the supply voltage Vbat.

[0006] A current limiter, CURRENT LIMITER, is normally present for limiting the current Ic at a maximum value Icl to ensure the functioning of the device within the allowed temperature range. The current limiter is activated when the voltage drop at the nodes of a sensing resistance RSENS overcomes a certain value and it drains a current Ilim equal to the difference between the current Iref and a replica, scaled by a factor A1, of the driving current Ib that is equal to the ratio between the limitation current Icl at the prescribed working conditions and the gain of the power device. In these conditions, the power stage will work in the direct biasing region of its current characteristic, thus causing strong power dissipation because of the Joule effect.

[0007] Diagrams of the main signals as a function of the time and of the temperature are also depicted in FIG. 1B, at the left and at the right of the dashed line portion thereof, respectively. Should the turn on signal remain high because of any malfunction, starting a current limitation phase, the power stage will dissipate a far greater power than under nominal working conditions, thus increasing the temperature of the integrated circuit. When a certain pre-established temperature TSOFT is reached, the soft thermal shut down circuit, SOFT THERMAL SHUT DOWN, activates itself and absorbs a current Ith linearly increasing with the temperature with a coefficient K1.

[0008] The current variations in each block must satisfy the 1^(st) Kirchhoff's law at node A, that is: ${\frac{Ib}{A\quad 1} = {{Iref} - {Isum}}};\quad {{Isum} = {{I\quad \lim} + {Ith}}};\quad {{Iref} = {cost}};$

[0009] In the temperature range from TSOFT and TSTART, the current Ith increases of the same amount as Ilim decreases, thus keeping Isum constant, while the current amplifier “A1 _(I)” provides the drive current Ib that is necessary to force the required current Ic in the power integrated transistor.

[0010] On the contrary, for a temperature T>TSTART an increase of the current Ith implies an increase of Isum with a consequent decrease of the input current of the current amplifier A1 _(I). Therefore, a decrease of the current Ib and thus a decrease of the maximum current Icl that may flow in the power transistor is obtained. As it is possible to note, there is a temperature TSTOP at which the current Icl(T) is zero even if a turn on signal IN at a high logic level is present. As a matter of fact, should the turn on signal be always high, the temperature TSTOP would be never attained. This is so because the circuit will eventually reach a temperature lower than TSTOP, at which point the power dissipated in the circuit equals the rate of heat dissipation in the environment.

[0011] The temperature TSTOP should be lower than the maximum junction temperature that may be tolerated by the integrated power transistor and/or be lower than the temperature at which unacceptable variations of the bandgap voltage, customarily used by the circuit as reference voltage, would take place. Preferably the temperature TSTOP is lower than 190° C. and the temperature TSTART, which is prescribed by specifications, is not lower than 150° C.

[0012] The difference ΔT=TSTOP−TSTART determines the value of the coefficient K1 of the soft thermal shut down circuit. It is not possible to set a relatively low TSTOP, i.e. close to TSTART, by setting a certain value K1, because of the problem of global stability of the system. In fact, should K1 be too great there would be abrupt variations of Ib with temperature. This would cause a consequent undesired oscillation of the output voltage Vc in proximity of the thermal equilibrium temperature. For this reason the values of K1 should be limited such to establish a TSTOP preferably of about 180° C. to 190° C.

[0013] The circuit of FIG. 1A may be improved by using an auxiliary thermal sensor TON THERMAL SENSOR, as depicted in FIG. 2A. Diagrams of the main signals as a function of the time and of the temperature are also depicted in FIG. 2B. The auxiliary thermal sensor enables the soft thermal shut down circuit when a certain temperature TON has been overcome by the integrated circuit. In this way, the soft thermal shut down circuit does not interfere in the normal functioning of the device if the circuit temperature is lower than the activation temperature.

[0014] These well known approaches do not fully address the problem of the instability of Vc. At best, the oscillations of the collector voltage Vc are limited in order to prevent inducing overvoltages that may produce sparks at the plugs in the secondary circuit of the coil. For example, the detected functioning of the commercially available device VB025 of STMicroelectronics, a functional diagram of which is depicted in FIG. 2A, after a turn on pulse IN (Ch1) lasting relatively for a long time (80 seconds), is illustrated in FIG. 8. Evident oscillations of the collector voltage (Ch3) can be noticed, when the collector current (Ch4, 2A/div), that is the current circulating in the primary circuit of the coil, diminishes.

SUMMARY OF THE INVENTION

[0015] It has been found and is the object of the present invention to provide a thermal control circuit and a related method of soft thermal shut down of an integrated power transistor that reduces the amplitude of the oscillations of the collector voltage of the power transistor. This is obtained by employing a current amplifier A1(T)_(I) having a variable gain that is controlled by the SOFT THERMAL SHUT DOWN circuit. The invention is directed to reducing the gain of the amplifier when the temperature increases, instead of reducing the current provided to the amplifier through the soft thermal shut down circuit, as in the known devices. In this manner, the value of Ib is reduced, but differently from the circuit of the prior art. At the same time also its oscillations, due for example to the input noise of the amplifier, are reduced.

[0016] More precisely, the thermal control circuit is for an integrated power transistor and comprises a current generator controlled by a turn on signal, a sensing resistance in series with the power transistor, a current limiter enabled when the voltage drop at the nodes of the sensing resistance overcomes a certain value, and a current amplifier coupled to the output node of the controlled current generator producing a drive current that is injected on the control node of the power transistor. The thermal control circuit may also include a soft thermal shut down circuit whose state of conduction increases as the temperature increases thereby progressively reducing the drive current of the power transistor.

[0017] The circuit of the invention controls the voltage on the power transistor in a significantly more effective manner than the known circuits because the current amplifier has a gain that varies as a function of the state of conduction of the soft thermal shut down circuit, and, therefore, as a function of the temperature.

[0018] A further object of the invention is to provide a method of soft thermal shut down of a power transistor that allows a reduction of the oscillations of the collector voltage. This method, implemented with a thermal control circuit of the invention, substantially includes reducing progressively the gain of the drive current amplifier as the temperature increases until a thermal equilibrium is reached.

BRIEF DESCRIPTION OF THE DRAWINGS

[0019] The different aspects and advantages of the invention will appear even more evident through a detailed description of several embodiments and by referring to the attached drawings in which:

[0020]FIGS. 1A to 2B are known thermal control circuits of a power transistor, as in the prior art;

[0021]FIGS. 3A to 4B are two possible embodiments of the circuit of the invention;

[0022]FIG. 5 is a schematic diagram of an embodiment of the variable gain amplifier of the circuits of FIGS. 3A to 4B;

[0023]FIG. 6 is a graph of plots of the main signals of the prior art circuit as shown in FIG. 1;

[0024]FIG. 7 is a graph of plots of the main signals of the circuit of the invention as shown in FIG. 3;

[0025]FIG. 8 are traces of results of a simulation of the functioning of the prior art circuit as shown in FIG. 2A.;

[0026]FIG. 9 are traces of results of a simulation of the functioning of the circuit of the invention as shown in FIG. 4.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0027] The thermal control circuit of the invention differs from known circuits by employing a current amplifier whose gain varies in function of the temperature. This effectively compensates for the fact that oscillations of the output voltage Vc are principally due to the variations on temperature of the drive current of the power device Ib upon the changing of the temperature when the output current Ic is relatively small because of the intervention of the soft thermal shut down circuit, SOFT THERMAL SHUT DOWN. Such generally large oscillations are effectively attenuated by employing a drive current amplifier whose gain becomes smaller upon the raising of the temperature at higher temperatures.

[0028] The diagrams of two different embodiments of the circuit of the invention are depicted in FIGS. 3A and 4A. As understood with additional reference to the diagrams of FIGS. 3B and 4B, the variation of Ib is not linear in the temperature range between TSTART and TSTOP, as for the prior art circuits of FIGS. 1A and 2A. Instead, the current decreases exponentially. Therefore, there is a rapid reduction of Ib and thus a rapid reduction of the current flowing in the power transistor. Such a characteristic is due to the fact that the gain substantially decreases as does the current Ic, and thus as much as the current Ib does. In fact the gain of the current amplifier is relatively high at low temperatures, that is, at the turn on of the power transistor when the current Ic is large. The gain progressively drops as the temperature increases, and while the transistor attains its steady state temperature and is crossed by a smaller current Ic.

[0029] The diagrams of FIGS. 3B and 4B, as well as those of FIGS. 1B and 2B, actually show in a simplified manner the true signals. In particular the variation of Ic(T) between the temperatures TSTART and TSTOP is depicted as being linear for sake of simplicity, but in general it is not linear because of the variation of the gain HFE=Ic/Ib of the power stage in function of the current Ib.

[0030] By comparing the circuit of FIGS. 3A and 3B to the circuit of FIGS. 1A and 1B (or, similarly, the circuit of FIGS. 4A and 4B to the circuit of FIG. 1A and 1B), it may be noted that the thermal coefficient K2 of Ith of the soft thermal shut down circuit is different from that of FIGS. 1A and 1B. Because of the particular embodiment of the block A1(T)_(I), it is possible to have larger coefficients K2 than in the known prior art circuits. Therefore, it is possible to obtain a lower temperature difference ΔT=TSTOP−TSTART than the corresponding temperature difference of the prior art circuits without being affected by the above mentioned problems of instability of the output voltage Vc.

[0031] A preferred embodiment of the controlled gain current amplifier A1(T)_(I) is depicted in FIG. 5. The shown amplifier is realized with BJTs, but it may alternatively be realized with MOS transistors instead of BJTs, as well as in an alternate form with transistors having a conductivity of opposite kind to that shown in the figure. The amplifier includes a plurality of stages. The first stage is a current mirror comprising transistors T6, T7 and emitter resistances R7, R8. Of course, this stage may be omitted if the current generator Iref, driven by the turn on signal IN, is coupled to ground instead of to the supply voltage.

[0032] The second stage is provided by another current mirror, preferably with recovery of the base current realized by the transistors T1, T2, T5 and having emitter degeneration resistances R1, R2. Finally a third emitter degenerated output current mirror is provided by transistors T3, T4 and degeneration resistances R3 and R4.

[0033] As it is well known, components R5, R6 and T5 serve only to improve the precision of the current mirror provided by transistors T1, T2. By straightforward calculations one may find that the value of the collector current circulating in trans T2 is given by the following formula: $\begin{matrix} {{{Ic}_{({T\quad 2})} = {\frac{1}{R_{2}} \cdot \left\lbrack {{{Iref} \cdot R_{1}} + {V_{T} \cdot {\ln \left( \frac{A_{2}}{A_{1}} \right)} \cdot \left( \frac{Iref}{{Ic}_{({T\quad 2})}} \right)}} \right\rbrack}}\left( {{{valid}\quad {for}\quad {Ith}} = 0} \right)} & (1) \end{matrix}$

[0034] where A1 and A2 are the respective areas of transistors T1 and T2, and VT is the thermal voltage. Generally the amplifier is realized such to make the first term, between the square brackets, much greater than the second term, thus rendering the latter negligible and the ratio between the currents Ic(T2) and Iref dependent only from the ratio of the resistances R1 and R2: $\begin{matrix} {{Ic}_{({T\quad 2})} \approx {\frac{R_{1}}{R_{2}} \cdot {{Iref}\left( {{{valid}\quad {for}\quad {Ith}} = 0} \right)}}} & (2) \end{matrix}$

[0035] The current mirror provided by transistors T1, T2 does not necessarily need base current recovery, thus eliminating devices R5, R6 and T5 and connecting the bases of transistors T1 and T2 to the collector of transistor T1. The opposite is true for the third current mirror. It may be a current mirror with base current recovery or, for sake of simplicity, even without emitter degeneration resistances R3 and R4. In this last case, the collector current of transistor T4 is amplified with respect to the collector current of transistor T3 not according to a function (1), but according to a different function as it well known in the art. Whichever may be the configuration of the variable gain amplifier, the resistances R1 and R2 must be present for reducing the amplification ratio of the current mirror provided by transistors T1, T2, simply by forcing a current Ith, that increases with temperature, through the soft thermal shut down circuit.

[0036] The difference between the invention and the prior art approaches may be appreciated from examining the comparative simulations of FIGS. 6 and 7. The simulation of a prior art circuit is shown in FIG. 6, while a comparable simulation carried out after appropriate assignation of the values of the components of the current of the invention of FIG. 5, is shown in FIG. 7. In both simulations the thermal sensor TON THERMAL SENSOR, which normally switches at 150° C., was disabled in order to show the behavior under the general conditions of FIGS. 1A, 1B, 3A and 3B. By enabling the sensor, there are instantaneous variations of Ith and Ilim starting from the temperature TON, as highlighted by the curves of FIGS. 2B and 4B.

[0037] The circuit of the invention of FIG. 4A was tested for verifying the performance improvements obtained, given that it was not possible to perform a transient simulation with relative temperature variation caused by thermal dissipation. Such a circuit was obtained by modifying the scheme of the connections of the known circuit of FIG. 2A and using the current amplifier of FIG. 5.

[0038] The experimental diagrams, depicted in FIG. 9, were obtained under the same test conditions that produced the results depicted in FIG. 8. As it may noticed, after the switching of the turn on signal IN (Ch1), there is a brief transient during which the collector current Ic (Ch4) diminishes, while the collector voltage (Ch2) reaches the steady state value. The comparison between the two figures shows clearly how, using the circuit of the invention, the collector voltage of the power transistor undergoes oscillations that are much smaller than the oscillations that occur in the prior art circuits. 

That which is claimed is:
 1. A thermal control circuit for an integrated power transistor comprising a current generator (Iref) controlled by a turn on signal (IN), a sensing resistance (RSENS) in series to said power transistor, a current limiter (CURRENT LIMITER) acting when the voltage drop on the sensing resistance (RSENS) overcomes a certain value, a current amplifier (A1 _(I)) coupled to the output node of said controlled current generator (Iref) outputting a drive current (Ib) that is injected on a control node of said power transistor, a soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) whose state of conduction is enhanced as the temperature increases for reducing the drive current of the power transistor, characterized in that said current amplifier (A1 _(I)) has a gain that varies in function of the state of conduction of said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN).
 2. The thermal control circuit of claim 1, characterized in that said current amplifier (A1 _(I)) is a multistage amplifier and said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) is functionally coupled to a signal node of an output stage of said multistage amplifier (A1 _(I)) and drains a current from said node in function of its state of conduction.
 3. The thermal control circuit according to one of claims 1 and 2, wherein said current amplifier (A1 _(I)) comprises at least a first current mirror (T1, T2) of a first type of conductivity, provided with emitter degeneration resistances (R1, R2), crossed by the bias current produced by said current generator (Iref) or by a scaled replica thereof and the current nodes of the output transistors (T2) of the current mirror are coupled respectively to said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) and to said current limiter (CURRENT LIMITER), and an output current mirror (T3, T4) in cascade to said first current mirror (T1, T2) and having a conductivity of opposite type, said amplifier (A1 _(I)) outputting said drive current (Ib) as the sum of the currents circulating in the transistors composing said output current mirror (T3, T4).
 4. The circuit of claim 3, wherein said amplifier comprises an input current mirror (T6, T7) having said opposite kind of conductivity and producing said scaled replica of the biasing current (Iref).
 5. The circuit according to one of the claims 3 and 4, wherein said first current mirror (T1, T2) is a current mirror with bias current recovery.
 6. The circuit according to one of claims from 3 to 5, wherein said output current mirror (T3, T4) and said input current mirror (T6, T7) are provided with emitter degeneration resistances (R3, R4; R7, R8).
 7. The circuit according to one of claims from 3 to 6, wherein said current amplifier (A1 _(I)) is made with BJT transistors.
 8. The circuit according to one of the claims from 1 to 7, further comprising a thermal sensor (TON THERMAL SENSOR) producing a signal for enabling or disabling said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) respectively when the temperature is greater or lower than a certain temperature (TON).
 9. The circuit according to claim 8, wherein said certain temperature (TON) is not lower than 150° C.
 10. The circuit according to any of the claims from 1 to 9, wherein said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) drains a current such to nullify the gain of said current amplifier (A1 _(I)) at a temperature not greater than 190° C.
 11. A method of soft thermal shutting down an integrated power transistor using a thermal control circuit comprising a current generator (Iref), controlled by a turn on signal (IN), a sensing resistance (RSENS) in series to said power transistor, a current limiter (CURRENT LIMITER), acting when the voltage drop on the sensing resistance (RSENS) overcomes a certain value, a current amplifier (A1 _(I)) coupled to the output node of said controlled current generator (Iref) outputting a drive current (Ib) that is injected on control node of said power transistor, a soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) whose conduction is enhanced as the temperature increases for reducing the drive current of the power transistor, the method comprising the steps of injecting said drive current (Ib) and activating said current limiter (CURRENT LIMITER) when a maximum current in the power transistor is reached and characterized in that it comprises varying the gain of said current amplifier (A1 _(I)) in function of the state of conduction of said soft thermal protection circuit (SOFT THERMAL SHUT DOWN).
 12. The method of claim 11, wherein said gain variation is carried out by attenuating the signal level in an output stage of said amplifier (A1 _(I)) by draining a current through said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) functionally coupled to a signal node of said output stage.
 13. The method according to one of claims 11 and 12, further comprising keeping disabled said soft thermal shut down circuit (SOFT THERMAL SHUT DOWN) by means of a temperature sensor, as long as the temperature does not exceed another pre-established value 